D5.3 Design of Sub-THz Transceiver Circuits and H-Band Plastic Waveguide

© COREnext 2023-2025 D5.3 Design of Sub-THz Transceiver Circuits and H-Band Plastic Waveguide D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 2 | 73 Revision v1.0 Work package WP5 Tasks T5.2 Dissemination level PU -Public Deliverable type R — Document, report (excluding periodic and final reports) Due date 30-11-2025 Submission date 28-11-2025 Deliverable lead CHAL Version v1.0 Authors Frida Strömbeck (CHAL), José Luis González Jiménez (CEA), Maciej Wojnowski (IFAG), Giuseppe de Astis (IFAT), Zulaicha Parastuty (IFAT), Samir Lagoug (IMS), Laurent Petit (RAD) Contributors All Task partners (see below) Reviewers Dick van den Broeke (NXP), Jonas Lindstrand (EAB), Florent Torres (EAB) Abstract This deliverable D5.3 describes the design and evaluation of sub-THz transceiver circuits, transceiver packaging, in-package interconnects, as well as polymer microwave fibre (PMF) and PMF transitions. It is based on the work done in task 5.2. These components will be the input for task 6.4, to enable energy efficient high data rate links in distributed data processing. Keywords PMF, eWLB, mm Wave transceivers, connector, D-band, H-band D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 3 | 73 Document Revision History Version Date Description of change Contributor(s) v0.1 23-10-2025 first draft v1.0 25-11-2025 submission version Frida Strömbeck (CHAL), José Luis González Jiménez (CEA), Maciej Wojnowski (IFAG), Giuseppe de Astis (IFAT), Zulaicha Parastuty (IFAT), Samir Lagoug (IMS), Laurent Petit (RAD) Contributing Partners Abbreviation Company name CHAL CHALMERS TEKNISKA HOGSKOLA CEA COMMISSARIAT AL ENERGIE ATOMIQUE ET AUX ENERGIES ALTERNATIVES EAB ERICSSON IFAG INFINEON TECHNOLOGIES AG IMEC INTERUNIVERSITAIR MICRO-ELECTRONICA CENTRUM NXP NXP SEMICONDUCTORS RAD RADIALL IMS INSTITUT POLYTECHNIQUE DE BORDEAUX NOK NOKIA NETWORKS GERMANY NNF NOKIA NETWORKS FRANCE IFAT INFINEON TECHNOLOGIES Disclaimer The information, documentation and figures available in this deliverable are provided by the COREnext project’s consortium under EC grant agreement 101092598 and do not necessarily reflect the views of the European Commission. The European Commission is not liable for any use that may be made of the information contained herein. Copyright Notice ©COREnext 2023-2025 D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 4 | 73 Executive Summary This deliverable consists of the work done in WP5 task 5.2. It focuses on the sub-THz link design to enable high data rate short range communication within the COREnext project. The work is divided into three different parts: transceiver design, in-package interconnects and polymer microwave fibre (PMF) design and characterization. The transceivers designed in the project are implemented in two different technologies. Infineon’s 90nm & 130nm silicon germanium (SiGe) BiCMOS process and Global foundries’ 45nm RF-SOI CMOS. The SiGe BiCMOS designs are optimized for wideband performance (one channel), while the CMOS design has a more complex baseband structure with multiple channels. Next part describes one of the packaging solutions, embedded wafer level ball grid array (eWLB), which was used to package samples of the SiGe BiCMOS circuits. An in-package PMF coupler based on a Vivaldi antenna is described and evaluated. The PMF shape and material is simulated and measured, and effects such as twisting is investigated. Two PMF transition solutions are developed, one to rectangular waveguide and one to the package. All designs of circuits, PMFs and couplers are used as base for the link demonstrations in WP6, task 6.4. D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 5 | 73 Table of Contents 1 Design and Characterization of Sub-THz Transceiver Circuits ................................................... 13 1.1 D-band Transceivers ........................................................................................................................................................... 13 1.1.1 D-band Transceivers in SiGe BiCMOS ............................................................................................................. 13 1.1.2 D-Band Transceivers in 45nm RF SOI CMOS ............................................................................................. 23 1.2 H-band Transceivers in SiGe BiCMOS ........................................................................................................................ 27 1.2.1 Lower H-band Tx/Rx in B11HFC ........................................................................................................................ 27 1.2.2 Lower H-band Tx/Rx in B12HFC ....................................................................................................................... 31 1.2.3 Upper H-band SiGe BiCMOS Transmitter in B12HFC ............................................................................. 35 2 Design and Characterization of In-Package Interconnect Elements ........................................ 40 2.1 PMF Coupler Concept ...................................................................................................................................................... 40 2.2 PMF Coupler Design and Analysis ................................................................................................................................ 42 2.2.1 Balun Design .............................................................................................................................................................. 44 2.2.2 Higher-Order Modes .............................................................................................................................................. 45 2.2.3 PMF Holder Design ................................................................................................................................................. 46 2.2.4 PMF Coupler Loss Analysis and Reduction ................................................................................................... 46 2.3 eWLB Package Variants .................................................................................................................................................... 48 2.3.1 RX and RX Packages for Active Characterization ....................................................................................... 48 2.3.2 Back-to-back Test Packages for Passive Characterization ..................................................................... 49 3 PMF and Waveguide Interconnect Design and Characterization ............................................... 51 3.1 D-Band PMF Transmission .............................................................................................................................................. 51 3.1.1 Twisting Effect on PMF .......................................................................................................................................... 51 3.1.2 Measurement Setup ............................................................................................................................................... 55 3.1.3 Transition and Insertion Loss (IL) ....................................................................................................................... 57 3.1.4 Attenuation Through PMF .................................................................................................................................... 57 3.2 D-Band Connector Development and Characterization .................................................................................... 58 3.2.1 Objective and Context .......................................................................................................................................... 58 3.2.2 Connector Architecture and Mechanical Design ........................................................................................ 58 3.2.3 RF Performance and Simulation Validation ................................................................................................ 60 3.2.4 Integration Strategy and Modularity ............................................................................................................... 63 3.2.5 Next Steps in Development ............................................................................................................................... 63 D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 6 | 73 3.2.6 H-Band Transposing (Discussion) ..................................................................................................................... 64 3.3 D-Band IC to Waveguide Connectivity ................................................................................................................ 65 3.3.1 Introduction ............................................................................................................................................................... 65 3.3.2 AFSIW Diplexer ........................................................................................................................................................ 66 3.4 H-Band Solutions .......................................................................................................................................................... 69 3.4.1 Observations and Performance ........................................................................................................................ 69 4 Conclusions ........................................................................................................................................... 70 5 References .............................................................................................................................................. 71 D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 7 | 73 List of Figures Figure 1: The metal stack of the B11HFC process .................................................................................................. 13 Figure 2: A block diagram of the D-band Tx/Rx in B11HCF .............................................................................. 14 Figure 3: Layout of the first version of D-band Tx/Rx in B11HFC .................................................................. 14 Figure 4: Schematic of the frequency doublers used in the Tx/Rx ............................................................... 15 Figure 5: Simplified schematic of the 6-stage common emitter amplifier ................................................ 15 Figure 6: Schematic of the up-converter mixer .................................................................................................... 16 Figure 7: Schematic of the down-converter mixer .............................................................................................. 16 Figure 8: Measured conversion gain for the first version of the D-band transmitter in B11HFC (IF= 4 GHz) ......................................................................................................................................................................................... 17 Figure 9: Measured saturated output power ........................................................................................................... 18 Figure 10: Conversion gain and output power for different input powers ................................................. 18 Figure 11: Measured conversion as a function of IF for the transmitter ....................................................... 18 Figure 12: Measured conversion gain for the first version of the D-band receiver in B11HFC (IF= 4GHz) ...................................................................................................................................................................................... 19 Figure 13: Measured conversion gain as a function of IF for the Receiver .................................................. 19 Figure 14: Captured eye diagram and I/Q constellation for a 28 GBd QPSK signal ............................. 20 Figure 15: Captured 34 Gbd 8-PSK signal .............................................................................................................. 20 Figure 16: Layout of the second version of D-band Tx/Rx in B11HFC ......................................................... 21 Figure 17: Amplifier topology used in the receiver ............................................................................................... 21 Figure 18: Measured conversion gain for the transmitter ................................................................................. 22 Figure 19: Measured saturated output power for the transmitter ................................................................. 22 Figure 20: Measured conversion gain for the receiver ...................................................................................... 22 Figure 21: Received QPSK constellation for a 24 Gbd signal ........................................................................... 23 Figure 22: Received 20 Gbd QAM-16 constellation for the two-meter link ............................................ 23 Figure 23: Main characteristics of the GF45nm RF SOI CMOS technology. ............................................ 23 Figure 24: Channel aggregation D-band transceiver architecture. ............................................................... 24 Figure 25: Block diagram of the D-band CMOS TX. ........................................................................................... 24 Figure 26: Photo of the two ICs of the D-band CMOS TX. .............................................................................. 25 Figure 27: Two channel version of the D-band CMOS RX. .............................................................................. 26 Figure 28: Two and four channels D-band CMOS RX ICs. ............................................................................... 26 D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 8 | 73 Figure 29: IF to BB multi-channel down-converter IC. ..................................................................................... 27 Figure 30: Block diagram for the H/Y-band Tx/Rx in B11HFC ...................................................................... 27 Figure 31: Layout of the lower H-band Tx and Rx in B11HFC .......................................................................... 28 Figure 32: The topology used in the H-band amplifier in B11HFC ................................................................ 28 Figure 33: Simplified schematic of the sub-harmonically pumped Gilbert cell up-converter mixer .................................................................................................................................................................................................. 28 Figure 34: Simplified schematic of the sub-harmonically pumped down converter mixer .............. 29 Figure 35: Measured conversion gain (Tx) for different LO power levels (IF= 4 GHz) .......................... 29 Figure 36: Measured output power (Tx) as a function of input power (RF = 136 GHz, IF = 4 GHz) 30 Figure 37: Measured conversion gain as a function of IF frequency. LO carrier = 240 GHz ............. 30 Figure 38: Measured conversion gain for the receiver ...................................................................................... 30 Figure 39: Measured conversion gain for the Rx as a function of IF frequency (LO carrier = 240 GHz) ................................................................................................................................................................................................... 31 Figure 40: Setup for a one-meter long PMF link using the B11HFC Tx/Rx at 237 GHz ....................... 31 Figure 41: The stack in Infineon B12HFC process .................................................................................................. 31 Figure 42: Block diagram of the lower H-band Tx in B12HFC ........................................................................ 32 Figure 43: Layout of the fabricated Tx (lower H-band B12HFC) .................................................................... 32 Figure 44: Measured conversion gain for the Tx (IF=4 GHz) ........................................................................... 33 Figure 45: Measured saturated output power (IF=4 GHz) ................................................................................ 33 Figure 46: Measured conversion gain (LO = 240 GHz) ...................................................................................... 33 Figure 47: Measured output power (IF= 4 GHz and RF= 230 GHz) .............................................................. 34 Figure 48: Measured conversion gain (IF= 4 GHz and RF= 230 GHz) ......................................................... 34 Figure 49: Block diagram of the lower H-band Rx in B12HFC ........................................................................ 34 Figure 50: Layout of the fabricated Rx (lower H-band B12HFC) .................................................................. 34 Figure 51: Measured conversion gain (IF= 4 GHz) ................................................................................................ 35 Figure 52: Measured conversion gain (LO=240 GHz) ........................................................................................ 35 Figure 53: Block diagram of the upper H-band transmitter ............................................................................ 35 Figure 54: Schematic of the H-band transmitter ................................................................................................ 36 Figure 55: A photo of the fabricated circuit ............................................................................................................ 37 Figure 56: Measured conversion gain for different RF frequencies (IF=1 GHz) ....................................... 37 Figure 57: IF sweep using an LO of 263 GHz ......................................................................................................... 37 Figure 58: Measured saturated output power for the transmitter ................................................................ 38 D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 9 | 73 Figure 59: Measured output power as a function of input power levels .................................................... 38 Figure 60: Measurement setup during the transmission test ......................................................................... 38 Figure 61: 32 Gbps OOK transmission eye diagram ............................................................................................ 39 Figure 62: 40 Gbps QPSK transmission using an IF carrier of 16 GHz ........................................................ 39 Figure 63: Measured output spectrum at 280 GHz and 4 Gbps QPSK modulation ............................ 39 Figure 64: PMF coupler realization (a) on PCB and (b) in package ............................................................... 40 Figure 65: Embedded Wafer Level Ball Grid Array (eWLB) packaging technology ................................ 41 Figure 66: Concept of Vivaldi antenna in eWLB for PMF coupler ................................................................ 42 Figure 67: (a) 3D simulation model of the PMF coupler using Vivaldi antenna and (b) EM field distribution at package-PMF interface ..................................................................................................................... 43 Figure 68: Simulated S-parameters of the PMF coupler using Vivaldi antenna .................................... 43 Figure 69: 3D simulation model of the balun in the RDL of eWLB ............................................................. 44 Figure 70: Simulated S-parameters of the balun in the RDL of eWLB ...................................................... 44 Figure 71: Simulated S-parameters of the PMF coupler using Vivaldi antenna with balun ............... 45 Figure 72: Simulated coupling losses of the fundamental mode and higher-order modes of the PMF coupler using Vivaldi antenna with balun ............................................................................................................... 46 Figure 73: Manufactured test boards (a) without and (b) with a PMF holder attached to the test board .................................................................................................................................................................................................. 46 Figure 74: Manufactured packages with PMF couplers with integrated RX and TX MMICs for active .................................................................................................................................................................................................. 48 Figure 75: 3D model of the manufactured package with PMF couplers with integrated TX MMICs for active characterization ............................................................................................................................................. 49 Figure 76: Manufactured test packages with back-to-back PMF couplers for passive characterization .................................................................................................................................................................................................. 49 Figure 77: Manufactured test boards for back-to-back PMF couplers for passive characterization (a) without and (b) with a PMF holder attached ......................................................................................................... 50 Figure 78: 3D views of the PMF with their transitions to circular metallic waveguide (a) without and (b) with a 𝜃 twist ................................................................................................................................................................. 52 Figure 79: PMF to circular waveguide transition used in experiments and simulations: (a) front view and (b) cross-section view showing the 5 sections of the transition ........................................................... 52 Figure 80: Theoretical, measured and simulated ∆𝑆21𝐻𝐸11𝑜𝑢𝑝𝑢𝑡𝑉 of the X-shaped PMF versus twist angle. ........................................................................................................................................................................... 53 Figure 81: Setup measurement using TDS method, left. E-field for different twist angles. .............. 54 Figure 82: Variation of the E-field for the square PMF, left. Variation of the E-field for a rectangular PMF with a ratio of 2 between width and height, right. ..................................................................................... 55 D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 16 | 73 Figure 6: Schematic of the up-converter mixer Figure 7: Schematic of the down-converter mixer The integrated circuits were evaluated in frequency domain using a Keysight PNA-X (67 GHz N5247A) used together with a VDI D-band extender WR 6.5 at the output of the Tx and as the input for the Rx. The LO was provided by a Keysight signal generator (Agilent 67 GHz PSG E8257D) which was synchronized to the PNA-X. The conversion gain (Figure 8) for the transmitter was measured with an input IF power of -15 dBm and an LO power fed into the quadrupler of 4 dBm. The IF was a fixed sinusoidal at 4 GHz and both D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 17 | 73 the lower sideband (LSB) and the upper sideband (USB) was measured sweeping the RF frequency between 110 - 170 GHz. The single-ended IF input was split into the four channels using a commercial hybrid (Marki QH-0440) and two baluns (Marki BAL-0050). The quadrature-phase IF signals were further connected to the GSSGSSG probe through external DC blocks. Figure 8: Measured conversion gain for the first version of the D-band transmitter in B11HFC (IF= 4 GHz) The difference in signal strength between the different sidebands is known as sideband suppression. Sideband suppression is a measure of I/Q (im-) balance. I/Q imbalance is a result of either an amplitude difference between I and Q, or a phase difference that is deviating from the ideal 90 degrees. The result is a distortion of the signal and can be measured in a signal-todistortion ratio. SDR= 10log61+ϵ! "+ϵ! "tan"(Δ ϕ!) ϵ! "+ tan"(Δϕ!)@ where εR is the amplitude error and ΔΦR is the phase error. For an SDR above 20 dB the phase error has to be lower than 6 degrees or the amplitude error less than 10 % [2]. Known distortion can be dealt with, but it is increasingly difficult to implement at high data rates, in real-time, which is why it can be seen as noise in those cases. The 3-dB RF bandwidth measured was between 125-165 GHz, corresponding to 40 GHz. The sideband suppression was approximately 20 dB, with a maximum of 25 dB at 152 GHz. The saturated output power was measured (Figure 9), and the peak output power was 0dBm. D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 18 | 73 Figure 9: Measured saturated output power An input power sweep was made between -25 and 5 dBm, and the corresponding output power and conversion gain can be seen in Figure 10. It can be seen that the transmitter starts to go into compression for higher power levels than -15 dBm. Figure 10: Conversion gain and output power for different input powers The conversion gain as a function of IF frequency was evaluated as well (Figure 11). The measurement was done from 4 GHz as this is the lowest frequency the external hybrid is working for. The estimated 3-dB bandwidth is 18 GHz. Figure 11: Measured conversion as a function of IF for the transmitter D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 19 | 73 The receiver was evaluated in the same way as the transmitter. The conversion gain as a function of RF frequency is shown in Figure 12. The Rx was measured to have the same 3-dB bandwidth as the Tx (125-165 GHz). Figure 12: Measured conversion gain for the first version of the D-band receiver in B11HFC (IF= 4GHz) The IF bandwidth for the receiver (Figure 13) was measured to be less than the IF bandwidth for the transmitter, indicating that the receiver is the limiting factor for a transceiver link using these circuits. A change in topology of the receiver, and/or adding IF amplifier/equalizer could potentially improve the performance. Figure 13: Measured conversion gain as a function of IF for the Receiver The Tx and Rx were evaluated in a one-meter PMF link. The data input was provided by a Keysight M8195A arbitrary waveform generator (AWG), where a pseudorandom binary sequence (PRBS-10) was generated using root raised cosine pulse shaping with a roll off of 0.7. Direct I/Q modulation was used during the link measurements. The LO was provided by a Keysight signal generator (Agilent 67 GHz PSG E8257D), which was shared by Tx and Rx using a power splitter. A Teledyne D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 20 | 73 LeCroy LabMaster 10-100Zi was used to capture the output signal from the Rx. 56 Gbps were reached error free (BER<10-12) using QPSK modulation (Figure 14). Figure 14: Captured eye diagram and I/Q constellation for a 28 GBd QPSK signal Highest data rate achieved was 102 Gbps using 8-PSK modulation (Figure 15). The error rate in this case was 2.1x10-3. Figure 15: Captured 34 Gbd 8-PSK signal For more information see publication (A Beyond 100 Gbps Polymer Microwave Fiber Communication Link at D-band, F. Strömbeck, Y. Yan and H. Zirath, 2023) [3]. 1.1.1.2 Second Version Tx/Rx in B11HFC The second version of the D-band Tx/Rx in SiGe BiCMOS can is shown in Fehler! Verweisquelle konnte nicht gefunden werden.. The size of the circuit was 0.7mm x 3 mm like the first version. The Tx was the same as the first version except for few minor changes. The passive elements (Marchand baluns and 90-degree hybrid) were modified to achieve a larger sideband suppression, and the upconverter mixer included a common emitter stage at the output as a buffer amplifier stage to increase the output power. D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 21 | 73 Figure 16: Layout of the second version of D-band Tx/Rx in B11HFC The receiver included a change in passive elements (as the transmitter) and uses another amplifier topology (Figure 17). The amplifier design is a differential cascoded design using two-stages. This design was used to achieve higher gain compared to the earlier amplifier topology. Figure 17: Amplifier topology used in the receiver Both circuits were evaluated using the same setup as the previous pair (Chapter 1.1.1.1). The conversion gain (Figure 18) and the saturated output power (Figure 19) was improved compared to the first version. The RF bandwidth was shifted down approximately 5 GHz. Unfortunately, no improvement in the sideband suppression was achieved. D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 22 | 73 Figure 18: Measured conversion gain for the transmitter Figure 19: Measured saturated output power for the transmitter For the receiver, there was also an improvement in the conversion gain (Figure 20). The RFbandwidth is flat between 125-170 GHz, and the sideband suppression is unchanged compared to the first version. Figure 20: Measured conversion gain for the receiver Since the largest benefit of the circuits in the second version is the improved gain and output power, longer PMF links were tried. A two-meter PMF link was tested, and 48 Gbps were reached error free. The constellation can be seen in Figure 21. D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 23 | 73 Figure 21: Received QPSK constellation for a 24 Gbd signal The highest data rate reached over this two-meter link was 20 Gbd QAM-16 transmission (Figure 22), corresponding to 80 Gbps, which had a bit error rate of 8x10-4. Figure 22: Received 20 Gbd QAM-16 constellation for the two-meter link For more information see publication (An 80 Gbps QAM-16 PMF Link Using a 130 nm SiGe BiCMOS Process, F. Strömbeck, Y. Yan and H. Zirath, 2023) [4]. 1.1.2 D-Band Transceivers in 45nm RF SOI CMOS Another technology used in the COREnext project is Global Foundries’ 45nm RF-SOI CMOS. This process features partially depleted SOI transistors and thick metal layers back-end-of-line (BEOL) allowing for high quality factor on-chip passives at mmW frequencies as shown in Figure 23. The figure also shows that this technology attains fmax values up to 310 GHz, which are well suitable for circuits operating at the D-band (110-170GHz). Figure 23: Main characteristics of the GF45nm RF SOI CMOS technology. The architecture chosen for the CMOS D-band transceivers is based on channel aggregation. The goal is to be able to cover a large RF band in order to support a high-data rate with moderately complex modulation schemes (such as 16-QAM) but not to impose this band at the digital frequency interfaces. This goal is achieved using multi-channel transceiver architectures that D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 24 | 73 combine narrower channels from the BB and IF bands into adjacent spectral locations at the Dband, such that the aggregated data send into each of the sub-channels total the aggregated highdata rate. In the example shown in Figure 24 a total of 16 channels each one occupying 2.16 GHz of BW totalize 34.56 GHz of band at D-band, allowing more than 100 Gb/s of data rate using single carrier (SC) 16-QAM modulation. Figure 24: Channel aggregation D-band transceiver architecture. 1.1.2.1 BB to D-band TX Chipset in CMOS 45nm RFSOI CMOS The transmitter circuit that provides the D-band signal to the transmitter end of the PMF is composed of two different ICs. The overall architecture for a reduced version composed of just 8 of the 16 channels shown in Figure 24 is shown in Figure 25. The first chip is a BB to IF channel aggregation up-converter. The second one is an IF to D-band multi-channel TX. Figure 25: Block diagram of the D-band CMOS TX. The BB-to-IF up-converter and channel-bonding IC is composed of four similar lanes, each one taking an I/Q differential BB input. Each lane includes a 2nd order filter, an I/Q up-conversion mixer ! " # A B C ' (FGGF+I-../MN B C ' ! 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More details can be found in [5] The IC shown in Figure 26, left. The IF-to-D-band up-conversion and transmission is implemented by a two-channel TX. Each of the two lanes is composed of an IF amplifier, an up-conversion mixer and a power amplifier (PA). In the fabricated IC prototype, both IF signals are taken from the same input connector and then split on-chip for simplicity. The LO signals for each lane are generated on-chip using the frequency multiplication technique presented in [6], with frequencies 82.08 GHz and 90.72 GHz, respectively. They are generated from a common input reference (REF IN) signal of 4.32 GHz generated using a commercial PLL. The D-band TX brings each of the IF input signals to adjacent sub-bands (LB and UB) centred at 143.34 GHz and 152.28 GHz, respectively, covering a frequency range of 17.28 GHz. These two signals are provided at separated outputs. The two outputs are combined on-board using filter and passive diplexer as will be explained in a further section. The IC shown in Figure 26, right. Figure 26: Photo of the two ICs of the D-band CMOS TX. 1.1.2.2 D-band Rx Chipset in CMOS 45nm RFSOI CMOS The RX takes D-band the signal from the PFM and brings it to BB at separate sub-channels. It is composed of a multichannel D-band RX IC that brings several sub-bands from the D-band range to the same IF band at separate outputs (see Figure 27 for a two sub-bands architecture). The IC photo for the two channels version is shown in Figure 28, left. A version with four channels has been also designed and fabricated, as shown in Figure 28, right. In both the 2-channel and the 4channel version each down-conversion lane is similar. They differ only on the passive circuits that are sized differently. Each LNA consists of four stages with capacitively neutralized pseudodifferential common source cells. The differential mode and the matching between the LNA stages are achieved with transformers. The mixer is a balanced Gilbert cell followed by a single-stage differential amplifier (IFA) centered at the IF band. The multiple required LOs are generated in the D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 32 | 73 Taking advantage of the higher ft/fmax of the B12HFC technology Tx and Rx versions using a fundamental mixer were designed. The block diagram of the transmitter (Figure 42), displays a new architecture of the LO chain. One potential issue using the fundamental mixer design is that it can be cumbersome to achieve enough LO drive power for the mixer core. A problem could be that the H-band LO buffer amplifier saturates at significantly low output power levels. To avoid this, the hybrid was moved before the buffer amplifier, splitting the input power to two buffer amplifiers instead. Furthermore, this Tx/Rx pair used balanced I/Q input to be able to use higher order inphase/quadrature modulation while still using direct modulation. Figure 42: Block diagram of the lower H-band Tx in B12HFC The fabricated Tx (Figure 43), measures 2.7mm x 0.95 mm. Figure 43: Layout of the fabricated Tx (lower H-band B12HFC) During the VNA measurements for these circuits (Tx/Rx), we used WR 4.3 extender. However, the probe used was a WR 3.4 meaning that some mismatch between the extender and probe occurred. Furthermore, the measurements below 220 GHz are compromised, since the probe only works properly from 220 GHz. Figure 44, shows the measurement result of the conversion gain. The sideband suppression is almost 20 dB, and the bandwidth is up to 240 GHz. The peak output power (Figure 45) was -4dBm. The IF frequency used was 4 GHz. D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 33 | 73 Figure 44: Measured conversion gain for the Tx (IF=4 GHz) Figure 45: Measured saturated output power (IF=4 GHz) The conversion gain using the IF frequency as a function can be seen in Figure 46. In Figure 47, the RF output power for different IF input power levels are shown. The signal starts to compress around -20 dBm IF power. Figure 46: Measured conversion gain (LO = 240 GHz) D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 34 | 73 Figure 47: Measured output power (IF= 4 GHz and RF= 230 GHz) Looking at Figure 48, it can be concluded that the LO multiplier chain doesn’t require more than – 10 dBm to saturate. Figure 48: Measured conversion gain (IF= 4 GHz and RF= 230 GHz) The receiver block diagram (Figure 49), and layout (Figure 50) are similar to the transmitter. Figure 49: Block diagram of the lower H-band Rx in B12HFC Figure 50: Layout of the fabricated Rx (lower H-band B12HFC) D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 35 | 73 The performance of the receiver conversion gain as a function of RF frequency (Figure 51), and as a function of IF frequency (Figure 52), shows a similar RF bandwidth as the Tx, while the IF bandwidth is a bit more limited. Figure 51: Measured conversion gain (IF= 4 GHz) Figure 52: Measured conversion gain (LO=240 GHz) This transceiver pair is planned to be evaluated through a link measurement. The link measurement will be performed on-wafer before the end of the project. 1.2.3 Upper H-band SiGe BiCMOS Transmitter in B12HFC A design in B12HFC aimed to cover the upper H-band was designed and fabricated in the June 2024 tapeout. The transmitter described in Figure 53 consists of a times eight LO frequency multiplier and a balanced up converter. Figure 53: Block diagram of the upper H-band transmitter The schematic (Figure 54) of the transmitter shows two cascaded frequency doublers using the same class B biased differential pair topology. The buffer amplifier (D-band) is a two-stage D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 36 | 73 common emitter configuration. The last-stage doubler does not have the cascoded transistor, due to limited gain at high frequencies. Finally, the mixer core is a balanced Gilbert cell. Figure 54: Schematic of the H-band transmitter The fabricated circuit, see layout Figure 55 uses many folded Marchand baluns, to convert single ended to differential, because of their broadband performance. The size of the circuit is 1590 μm x 630 μm including pads. D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 37 | 73 Figure 55: A photo of the fabricated circuit For the frequency domain measurements, a Keysight PNA-X (67 GHz N5247A) was used together with a VDI H-band extender WR 3.4 at the output. For a balanced IF input, an external balun (Marki BAL-0050) was used. The conversion gain for the transmitter was measured first by sweeping the RF frequency (see Figure 56Figure ), then the IF frequency was swept (Figure 57). Figure 56: Measured conversion gain for different RF frequencies (IF=1 GHz) Figure 57: IF sweep using an LO of 263 GHz The saturated output power (Figure 58) was measured for the transmitter, as well as the output power for different input power levels at 270 and 290 GHz RF (Figure 59). D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 38 | 73 Figure 58: Measured saturated output power for the transmitter Figure 59: Measured output power as a function of input power levels The setup (Figure 60) used during the transmission test, where the carrier was set to 300 GHz. Data input was provided by a Keysight M8195A arbitrary waveform generator (AWG), which generated a pseudorandom binary sequence (PRBS-11) using root-raised cosine pulse shaping with a roll-off of 0.8. Furthermore, direct conversion was used in the first part of the evaluation. The transmitted signal was captured through a probe that was connected to a WR 3.4 VDI Compact Converter (CC). The down-converted single-ended output signal from the CC was then connected to a Keysight UXR1104A Infinium Real-Time Oscilloscope for evaluation. The error vector magnitude (EVM), and SNR were measured. Figure 60: Measurement setup during the transmission test D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 39 | 73 The received eye diagram of a 32 Gbps transmitted on-off keying (OOK) signal is shown in Figure 61. The EVM was 11.8% and SNR=18.6 dB. The VDI compact converter has a conversion loss of 12dB, so in combination with the limited output power from the transmitter the signal is close to the detection limit of the oscilloscope. Figure 61: 32 Gbps OOK transmission eye diagram The transmitter was also evaluated using an IF carrier to achieve quadrature phase shift keying (QPSK) modulation transmissions. Symbol rates up to 20 GBd was tested, corresponding to 40 Gbps. The received I/Q constellation is displayed in Figure 62. The EVM=25.4% and SNR=11.9 dB. Figure 62: 40 Gbps QPSK transmission using an IF carrier of 16 GHz Finally, the output spectrum for a modulated QPSK signal (Figure 63) was captured. Figure 63: Measured output spectrum at 280 GHz and 4 Gbps QPSK modulation For more information see publication (A 0.3 THz Transmitter in 90-nm BiCMOS Technology for Energy-Efficient High Data Rate Communication, F. Strömbeck, Y. Yan and H. Zirath, 2025) [9]. D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 40 | 73 2 Design and Characterization of InPackage Interconnect Elements This chapter reports the design and characterization of the polymer microwave fiber (PMF) coupler realized in the embedded Wafer Level Ball Grid Array (eWLB) packaging technology. 2.1 PMF Coupler Concept There are two primary methods for coupling the RF signal into a PMF that is coupling via the RF board PCB) and coupling directly from the package. Each approach offers distinct advantages and challenges, particularly at high frequencies. The board-based coupling approach provides the advantage of scalability, enabling the design of larger couplers. It also offers significant flexibility in terms of placement and design, thanks to the typically large size and multilayer stack-up of RF boards. However, this method introduces additional challenges, such as the need for broadband interconnections between the chip and board or between the package and board at high frequencies. These interconnections can lead to increased losses and added tolerance, impacting performance. (a) (b) Figure 64: PMF coupler realization (a) on PCB and (b) in package Direct coupling from the package eliminates the need for board-level interconnections, offering a more compact and highly efficient solution. By integrating the coupler directly into the package, D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 41 | 73 the signal transition occurs with minimal losses and significantly fewer parasitic effects. This approach is particularly advantageous for high-frequency designs, as it removes the additional impedance mismatches and tolerances encountered with PCB traces and transitions. The performance degradation between the chip and board or between the package and board can still be acceptable for D-Band PMF couplers. However, at H-band frequencies, the loss and parasitic effects render the transitions highly inefficient, making in-package integration the only viable option. The packaging integration technology utilized is the eWLB [10], an advanced wafer-level packaging solution. This innovative platform extends the available area around the chip by providing a fanout region, enabling enhanced functionality and integration capabilities. eWLB leverages the use of low-loss materials along with high-resolution redistribution layers (RDL), which are key to its superior performance. These technological features make it possible to efficiently implement a variety of low-loss RF interconnections and antennas directly within the fan-out region. This capability is particularly advantageous for high-frequency applications, where minimizing signal loss and maintaining high performance are critical design priorities. Figure 65: Embedded Wafer Level Ball Grid Array (eWLB) packaging technology The proposed design for the PMF coupler integrated within the package utilizes a Vivaldi antenna, implemented on the RDL of the eWLB. This choice leverages the unique properties of the Vivaldi antenna and the advanced packaging capabilities of eWLB to meet the performance and cost requirements of high-frequency systems. D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 48 | 73 Table 4: Power loss distribution in standard and inverted PMF couplers assuming radiation loss reduced to zero The conducted loss analysis provides valuable insight into the individual loss contributions, guiding the direction of further PMF coupler improvements and indicating the potentially achievable performance. 2.3 eWLB Package Variants 2.3.1 RX and RX Packages for Active Characterization Figure 74 shows photographs of the manufactured eWLB packages with PMF couplers and RX and TX B11 MMICs for active characterization. In addition to the PMF coupler, the packages incorporate a very short single-ended LO interconnection to the PCB and a long, low-loss, matched differential transmission line for the IF. Figure 74: Manufactured packages with PMF couplers with integrated RX and TX MMICs for active Figure 75 shows the 3D simulation model of the manufactured TX package. !"#$$B&'( !"#$$B&)( !"#$ BCBD( F*H ,-$. BCB(F FFH L!M BCB(D F1H !"#$ BCBB( FH L!M BCBBF BH 2,P BCBB1 FH 4-$.#R BCB67 FBH L8. BCB7D 16H *+,-B./ 29W 2,P #;MW !012P$ D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 49 | 73 Figure 75: 3D model of the manufactured package with PMF couplers with integrated TX MMICs for active characterization 2.3.2 Back-to-back Test Packages for Passive Characterization Figure 76 shows the manufactured eWLB test packages featuring back-to-back PMF couplers for passive characterization. Figure 76 shows photographs of the manufactured test boards (a) without the PMF holder and (b) with the PMF holder attached. The PMF fiber protruding from the holder is clearly visible. Figure 76: Manufactured test packages with back-to-back PMF couplers for passive characterization D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 50 | 73 (a) (b) Figure 77: Manufactured test boards for back-to-back PMF couplers for passive characterization (a) without and (b) with a PMF holder attached D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 51 | 73 3 PMF and Waveguide Interconnect Design and Characterization In this chapter, the design (material and shape) of the PMF and PMF couplers is the focus. Solutions for both Dand H-band will be investigated. Furthermore, effects like twisting and bending are studied and evaluated. The two types of PMF connectors that COREnext has developed in the project are rectangular waveguide-to-PMF and directly from the package-to-PMF. 3.1 D-Band PMF Transmission The development of active technologies with smaller technological nodes has enabled the emergence of higher frequency transceivers, particularly in the D-band (110-170GHz) and H-band (230-325GHz). This increase in frequency raises a challenge for the passive component. Conventional wired connections (copper cable, optical fibres) are no longer suitable because they have too much loss. For applications ranging from 1 to 10 metres, plastic fibres seem to be more suitable and less lossy. 3.1.1 Twisting Effect on PMF PMF technology has been around since the 1970s but was reintroduced in 2011 [11]. However, as this fibre is not shielded, it leaves an evanescent field around the fibre. This raises the question of the robustness of this fibre in real-world conditions, i.e. when bent and twisted. Studies on bending have been carried out in recent years, concluding that the fibre has a maximum bending radius for each bandwidth [12]. It can be assumed that if the bending radius is several wavelengths, then the impact of this twisting can be neglected. However, no studies had been conducted on twisting along the fibre. A theoretical model was developed to understand this effect on the polarization for both fundamental hybrid 𝐻𝐸## $ and 𝐻𝐸## % modes (with V stand for vertical and H horizontal). D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 52 | 73 Figure 78: 3D views of the PMF with their transitions to circular metallic waveguide (a) without and (b) with a 𝜃 twist Figure 79: PMF to circular waveguide transition used in experiments and simulations: (a) front view and (b) cross-section view showing the 5 sections of the transition Table 5: Length and diameter of the transition sections shown in figure 63 Section I 1 2 3 4 5 Li(mm) 2.4 1.3 6 2.8 4 Di(mm) 8 7 6 5 4.2 The propagation modes in plastic waveguides are hybrid electric modes with the fundamental mode being 𝐻𝐸##. The propagation medium is assumed to be linear and homogeneous. With the previous assumptions, the Maxwell-Gauss equation can be written as follows: 𝐷 # # $ =𝜀![𝜀"]𝐸 # $ *(1) With 𝐷B B C the electric induction, 𝜀& the permittivity in the vacuum, [𝜀'] the second-order tensor describing the relative permittivity of the PMF and the 𝐸 # $ *electric excitation. Figure 78 above illustrates a straight and twisted plastic waveguide with its transitions to circular metallic waveguides. The transitions are not only used in experiments to interconnect laboratory instruments, but also in simulations to guarantee the proper excitation with a waveguide port of the plastic waveguide modes, as part of the electromagnetic wave propagates in the air surrounding the plastic waveguide. The first end of the PMF is fixed (port 1). Whereas the second end (port 2) is rotated by a θ twisting angle (as shown in Fig. 2) resulting in a continuous twist along the plastic waveguide. The PMF being subjected to a twist, equation (1) becomes function of the θ angle. In addition to the variation in relative permittivity tensor due to the θ angle, a rotation matrix with respect to the zaxis of angle θ must be introduced. This gives Equation (2): 𝐷(𝜃) B B B B B B B B B B C G=𝜀&𝑅((𝜃)[𝜀'(𝜃)]𝐸B C (𝜃)G(2) D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 53 | 73 Propagation takes place along the z-axis, so the relative permittivity tensor can be simplified. The coefficient will be invariant despite rotation through a θ angle. Hence: [𝜀'(𝜃)]=I𝜀'!!(𝜃)𝜀'!"(𝜃)0 𝜀'"!(𝜃)𝜀'"" (𝜃)0 0 0 𝜀'##JG (3) 3.1.1.1 Isotropic Case In the case of a fibre that has the particularity of being isotropic, the coefficients in the permittivity tensor will not change. Thus, the twist along the fibre will have no impact, i.e. the polarisation of the electromagnetic field will not follow the angle of twist. This result has been verified in simulation and measurement [13]. This result can be extended to fibres with a permittivity distribution that is almost constant in all directions (almost isotropic). This therefore applies to the cross-shaped guide, which is a circular plastic fibre with air in the middle, complemented by two rectangles at its centre. Thus, the variation for the 𝐻𝐸## $ mode transmission is given by the following equation: ∆L𝑆"#$%&&'()(* +L(𝜃)=10log#&(cos(𝜃)G𝐻𝐸##,-)(* $)(4) Measurement results are consistent with the theoretical and simulated results as shown in Figure 80 below. A 3 dB variation is obtained for a 45° twist angle, i.e., at the output port, half of the amplitude of the 𝐻𝐸##,*+,-. $ is transferred to 𝐻𝐸##,/-.,-. $, the other half being coupled to 𝐻𝐸##,/-.,-. %. Figure 80: Theoretical, measured and simulated ∆/𝑆#$ !"##$%&%' (/ of the X-shaped PMF versus twist angle. Despite a twist, the direction of the polarization does not rotate for the isotropic case. 3.1.1.2 Anisotropic Case The α angle is therefore the maximum angle such that 𝐸B C and 𝐷B B C fields are still collinear and the direction of polarization of the wave follows the twist angle (case 1). In case 2, the polarization D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 54 | 73 direction begins to deviate from the angle of twist. The receiver will then see a variation in ∆L𝑆"#$%&&'()(* +L(𝜃). The fields no longer follow the twist. The 𝐸B C field will no longer be colinear with the 𝐷B B C field. ⎩ ⎪ ⎨ ⎪ ⎧ G∆L𝑆"#$%&&'()(* +L(𝜃)=0GG𝑖𝑓G𝜃≤𝛼G𝑎𝑛𝑑Gtan𝜃≈𝜃G(𝑐𝑎𝑠𝑒G1) G∆L𝑆"#$%&&'()(* +L(𝜃)>0G𝑖𝑓G𝜃>𝛼G𝑎𝑛𝑑Gtan𝜃≠𝜃G(𝑐𝑎𝑠𝑒G2)(5) When the fibre is anisotropic, for example rectangular, the impact of the twist is visible. Indeed, if the ratio between the width and height is greater than 1.5, then the polarisation will follow the angle of twist. This result has been verified in simulation and measurement on an optical measurement bench using the TDS (Time Domain Spectroscopy) method. Rectangular fibres ranging from square fibres (width = height) to rectangular fibres with a width twice as large as the height were measured. Figure 81: Setup measurement using TDS method, left. E-field for different twist angles. Figure 81 shows the sub-THz measurement bench used to characterise the amplitude of the E field. This measurement bench uses TDS technology and allows measurements to be taken that provide an image of the E field from 50GHz to 10THz. Several rectangular waveguide shapes were measured to confirm the theoretical model developed. In order to facilitate 3D printing of plastic fibres, it was decided to vary only the width for a height of 0.8 mm. The height varies from 0.8 mm to 1.6 mm. The guide is positioned to maximise transmission in the fibre. The twist angle is controlled by a holder that rotates in its centre with graduations every degree. The Figure 81 shows the results for the rectangular guide with a width twice as large as the height. As the twist angle increases, the E-field amplitude decreases. This trend confirms the theoretical model. D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 55 | 73 Figure 82: Variation of the E-field for the square PMF, left. Variation of the E-field for a rectangular PMF with a ratio of 2 between width and height, right. Measurements were taken for the three fibres. Figure 82 shows the variation in amplitude at the resonance frequency of the fibres. Figure 82, left shows this difference for the square guide. Initially, the amplitude of the E field decreases up to 45°. When this angle is reached, only half of the E field passes through. Subsequently, the trend reverses. This is due to orthogonal polarisation, which becomes the polarisation most aligned with the receiver. Figure 82, right, shows the evolution of the amplitude of the E field as a function of the twist angle for the rectangular shape with a width twice as large as the height. In this case, theory explains that the field amplitude must be zero when the waveguide is rotated 90°. The figure confirms this trend. At 90°, less than 1% of the initial amplitude is measured at the output. This value is close to the measurement uncertainties and confirms the trend. This study on twist highlights the importance of choosing the right fibre shape when selecting an antenna or connector for reception. Depending on the shape, linear polarisation would be optimal if the fibre remains well aligned with the receiving antenna and the fibre is anisotropic (with a widthto-height ratio of 2). However, in other cases, circular polarisation would be more suitable. 3.1.2 Measurement Setup Two experimental campaigns were conducted jointly to evaluate the feasibility and performance of D-band polymer microwave fibres (PMFs) and associated waveguide transitions. These experiments took place in March 2024 (characterization of transmission lines) and June 2025 (characterization of connectors), as part of the COREnext project deliverables. The experimental setup was based on Rohde & Schwarz vector network analysers (VNAs) with Dband extenders, using both SOLT (TOSM) and TRL calibrations. Mechanical fixtures (photo on Figure 84Figure , below) included: • WR6.5 to circular metallic transitions, • Circular to PMF tapers, • PMF holders with centring foam and plastic rings, as shown hereafter D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 56 | 73 Figure 83: PMF to circular waveguide transitions Table 6: Dimension in mm of the transitions for (a) Hollow-core PMF (b) Circular PMF at D-band Section: 1 2 3 4 5 di(a) 1.86 1.9 2 2.2 2.4 Li(a) 1.7 1.5 1.6 1 1.1 di(b) 1.86 1.7 1.76 2.2 2.4 Li(b) 1.43 1.75 1.6 0.7 1.5 D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 57 | 73 Figure 84: The Measurement setup for D-band 3.1.3 Transition and Insertion Loss (IL) Different tapers were tested to connect WR6.5 rectangular waveguides to circular PMF-guided sections. Two types of PMF were considered: • 2.00 mm hollow PTFE fibres, and • 1.75 mm HDPE fibres (Zetamix-type). The insertion losses observed for back-to-back transitions (WR6.5 → circular → PMF taper) were typically ~1 dB in the D-band (110–170 GHz), with ~0.5 dB attributed to the WR6.5 to circular transition and ~0.5 dB to the PMF taper. Full through measurements (two transitions + PMF section) showed: • ILs between 1 and 2 dB, depending on taper quality and PMF diameter, • multiple reflections (S11, S22) due to mechanical inaccuracies and potential air gaps. These measurements showed good agreement with CST simulations and were consistent across several taper types (1.75 mm and 2.00 mm PMF). 3.1.4 Attenuation Through PMF Dedicated transmission measurements on PMF samples of 1 m, 2 m, and 4 m lengths allowed the extraction of the attenuation per meter α: • For the 2 mm PTFE hollow fibre, attenuation was in the range 3.9–5 dB/m between 110– 170 GHz, • Very good correlation with analytical attenuation models was found up to 150 GHz, as shown in the Figure 85, below. The ‘noisy’ behaviour observed beyond 150 GHz is confirmed by EM simulation analysis to be due to higher modes that start to be propagated beyond this frequency. These modes can be excited by small bendings of the fibre and are D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 64 | 73 Further developments will focus on industrialization aspects, including repeatability, reliability, and assembly procedures, as well as the extension to cryogenic-compatible versions for quantum applications. The development of PMF prototypes will focus on the incorporation of coating and sheath elements, alongside exploring materials and mechanical concepts that enhance secure fastening for hard materials. Adjustments will be made to accommodate both V-band and H-band specifications, with the adaptation to V-band being relatively straightforward. In contrast, the transition to H-band will present more significant challenges from an industrialization perspective. Furthermore, a new approach to PMFs will be elaborated, enhancing electromagnetic (EM) coupling mechanisms. This initiative aims to innovate more again in connector design while addressing the technical challenges associated with high-frequency RF applications. 3.2.6 H-Band Transposing (Discussion) Transposing the concept and design towards high-frequency connectors, such as those in the Hband, presents several notable challenges. The advancement of RF technology in sub-THz and above frequencies introduces critical constraints in connector design and performance. The need for extreme miniaturization leads to mechanical tolerances frequently below ±5 µm, making these connectors exceedingly sensitive to misalignment or gaps between mating components, which can severely affect their performance. Additionally, traditional metallic materials like brass and copper are reaching their performance limits, necessitating the use of surface finishes and metallization techniques, such as gold or silver plating, to maintain optimal conductivity. The impact of increased surface roughness significantly exacerbates issues related to conductivity at these frequencies. Moreover, the durability and repeatability of connectors raise concerns; mechanical fatigue, particularly in screw-on or push-on types, can result in performance drift over time and subject to temperature fluctuations. The variability in connectors can lead to significant challenges in maintaining reliable RF connections. Compounding these issues is the complexity of achieving a precise RF transition from waveguide to coaxial structure, where mode conversion and impedance matching become particularly difficult beyond 220 GHz. Finally, the limited industrial offerings and the absence of widely accepted interface standards further hinder the development and deployment of reliable connectors for these high-frequency applications. Addressing these challenges requires innovative solutions and rigorous testing to enhance the reliability and performance of RF connectors in high-frequency contexts. D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 65 | 73 Figure 94: H-Band PMF connector solution is valid in theory but raises important manufacturing challenges COREnext consortium has actively positioning itself throughout the entire interconnect chain for the D-band, which includes coaxial connectors, metallic and plastic waveguides, as well as hybrid assemblies. The expansion into H-band and sub-THz frequencies is paving the way for new applications such as ultra-high-speed communications exceeding 100 Gbps, high-resolution imaging, and radar systems, along with potential optical fiber replacement solutions for shortrange links. However, several technological bottlenecks remain unsolved. There is a need for mechanically robust and RF-optimized interfaces at frequencies above 220 GHz, along with precise characterization of dielectric and conductor losses near 300 GHz. Advanced packaging challenges related to thermal, mechanical, and RF co-design also necessitate focused attention. In line with this, COREnext consortium has a medium-term vision aimed at the industrialization of reliable, cost-effective H-band connectors and the establishment of a standardized sub-THz connector product line. This strategic direction will leverage ongoing developments to address the demands of high-frequency applications effectively. 3.3 D-Band IC to Waveguide Connectivity 3.3.1 Introduction This section presents the interconnection between the active part and the PMF fiber of a transmitter, the architecture of which is shown in the figure below. The active part of the CEA has two outputs. Each output has a different frequency, LB (Low Band) and UB (Upper Band). In order to combine both outputs, it is therefore necessary to use a D-band PCB diplexer. However, at present, the literature reports only one D-band diplexer compatible with PCB technology [24]. However, the losses are very significant. This section is devoted to the design of a new PCB diplexer with the least possible loss. D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 66 | 73 Figure 95: Transceiver architecture with the red part forming the diplexer 3.3.2 AFSIW Diplexer AFSIW (Air Filled Substrate Integrated Waveguide) technology has been in development since 2014 [25]. This multi-layer technology is unique in that it is filled with air in its center. These limits losses compared to other PCB technologies (SIW, microstrip, etc.). However, the PCB will inevitably be larger due to its lower relative permittivity. The system operates in the D band. Compactness is therefore no longer the main constraint for the passive part. In fact, the constraint is the ability to easily manufacture the circuit given that the wavelength is 2 mm. Thus, as AFSIW is a larger technology, it seems that it could be more easily manufactured using conventional PCB manufacturing processes. In order to have the most compact, manufacturable system with good RF performance, the decision was made to use dual-mode cavities. These cavities are larger than conventional cavities and therefore easier to manufacture. The second advantage is that dual-mode cavities create two poles and a transmission zero, which is necessary to comply with the specifications set at the beginning of the project. The bandpass filters were designed for the bandwidths of interest defined with CEA. A filter version with a wider bandwidth was designed and manufactured. D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 67 | 73 Figure 96: S-Parameter of the Low Band Filter Figure 96 shows the S parameters of the low-pass filter for operation with the CEA section. This filter is a 6-pole filter with 2 transmission zeros. One of the poles is at the top of the passband. In order to widen the rejection bandwidth, the transmission zeros are not placed in the same frequency. This also demonstrates the filter's reconfigurability. AFSIW is a technology for which there are currently only two articles on work using this technology at this frequency [26]-[27]. Only one of these papers presents a cavity filter, but with a circular and elliptical cavity with a shift in frequency between simulation and measurements [26]. Thus, in order to obtain results with good correlation between simulation and measurement, the prototypes were designed to be as robust as possible to variations in the manufacturing process. It should be noted that the filter does not include tuning to compensate for variations in the manufacturing process. Several filters in several passband bandwidths have been designed: • 138-147GHz • 150-158.5GHz • 140-160GHz All these filters use the same architecture but with different dimensions. In order to combine the two filters in a single output, a manifold is designed. The difficulty of the manifold results in the relative bandwidth of the system. It is also very important to note that the diplexer has a fractional guard band (1%) that is very small compared to what may exist in the literature (5-10%). This complicates the design of the manifold, which must respectively present an open circuit at f1 and then a closed circuit at f2 for channel 1 (and vice versa for channel 2). Where f1 is the center frequency of the Low Band (LB) and f2 is the center frequency of Upper Band (UB). D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 68 | 73 Figure 97: S-parameters of the Diplexer Figure 97 shows the S-parameters of the diplexer with all the mandatory interconnections (transitions to microstrip line, filters, manifold) for the final cointegration of the diplexer into the architecture shown in Figure 95. On the two bands of interest, insertion losses are in the range of 1 to 1.5dB. The isolation between the two channels is 19 dB in the worst case. A resonance at 162 GHz is visible. It is related to a resonance in the manifold. The first higher mode is at 173.5 GHz, i.e. 15 GHz above the frequency of interest. Finally, the matching in each channel is approximately 10dB. The project requires the ability to interface with waveguide connectors and integrated circuits. To address this aspect, transitions to its two outputs have been made. The diplexer was therefore manufactured so that it could be measured with a Vector Network Analyzer (VNA) using probes or waveguide output. A probe measurement technique for the diplexer was therefore developed in order to perform a rigorous multiport measurement. This measurement requires measuring all Sij with the third port matched to 50 Ohms, i.e. no reflection. A line allowing 20dB of loss to be added in order to achieve a good match was added. The measurements will be carried out at CEA before the end of the project. D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 69 | 73 Figure 98: Filter and Diplexer with probe and WR connector using AFSIW. 3.4 H-Band Solutions Initial feasibility tests for the H-band (220–330 GHz) were performed in March 2024 by RAD, CEA, and IMS using: • WR3.4 to circular waveguide transitions (micromachined), • PMF tapers to accommodate 0.889 mm PTFE fibres, and • Back-to-back and through configurations. 3.4.1 Observations and Performance • IL values in H-band tests ranged from 1.5 to 2 dB per transition pair. • The air-filled cavities around the tapers had a stronger influence on insertion loss and ripple, compared to D-band. • A multimode behaviour was suspected beyond ~300 GHz, particularly for the circular waveguides, as highlighted experimental overlays [8,15] RAD and partners provided WR3.4 transitions and PMF tapers to several COREnext partners for further testing and integration. D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 70 | 73 4 Conclusions This report presents the different parts designed and characterized in WP5 task 5.2. Solutions for two waveguide bands were developed. For D-band (110-170 GHz), the PMF loss was measured to be 3.9-5 dB per meter. The H-band (220-325 GHz) PMF is expected to have a significantly higher loss per meter, but that has not yet been verified. Multiple transceiver designs were developed and tested, using two different strategies. Single channel broadband designs were implemented in SiGe BiCMOS. A 16-channel design was implemented in RF SOI CMOS, requiring a more complex baseband structure. Both solutions were proven to be able to support the high data rate for D-band. For H-band transceivers SiGe BiCMOS shows promising results, but the designs need further development to reach their potential. A variety of PMF couplers were designed, covering both coupling from rectangular waveguide on PCB to PMF and directly from package to PMF. The benefit of using the waveguide interface is that it can be coupled with any waveguide component providing flexibility, but it requires an additional transition from package to PCB with associated losses. An in-package solution using Vivaldi antenna launcher was tested and looks promising. The work described in this report shows that all individual components achieve good performance that enables a successful execution of Task 6.4. D5.3 Design of sub-THz transceiver circuits and H-band plastic waveguide 71 | 73 5 References [1] R. Schmogrow et al., "Error Vector Magnitude as a Performance Measure for Advanced Modulation Formats," in IEEE Photonics Technology Letters, vol. 24, no. 1, pp. 61-63, Jan.1, 2012, doi: 10.1109/LPT.2011.2172405. [2] F. Horlin and A. 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