0.3-V Nanopower Biopotential Low-Pass Filter

Received June 12, 2020, accepted June 26, 2020, date of publication June 29, 2020, date of current version July 9, 2020. Digital Object Identifier 10.1109/ACCESS.2020.3005715 0.3-V Nanopower Biopotential Low-Pass Filter TOMASZ KULEJ 1, FABIAN KHATEB 2,3, AND MONTREE KUMNGERN 4 1Department of Electrical Engineering, Częstochowa University of Technology, 42-201 Częstochowa, Poland 2Department of Microelectronics, Brno University of Technology, 60190 Brno, Czech Republic 3Faculty of Biomedical Engineering, Czech Technical University in Prague, 3105 Kladno, Czech Republic 4Department of Telecommunications Engineering, Faculty of Engineering, King Mongkut’s Institute of Technology Ladkrabang, Bangkok 10520, Thailand Corresponding author: Fabian Khateb ([email protected].cz) This work was supported by the King Mongkut’s Institute of Technology Ladkrabang under Grant KREF026201, and in part by the National Sustainability Program under Grant LO1401. ABSTRACT This paper presents a compact power-efficient CMOS fourth-order low-pass filter suitable for electrocardiogram (ECG) acquisition systems. The CMOS structure of the proposed filter utilize the bulkdriven technique and operates in subthreshold region to achieve extremely low-voltage supply (0.3V) and nanopower consumption (0.676 nW) for cut-off frequency of 100 Hz. The filter was designed and simulated using 0.18 µm CMOS TSMC technology. The total input referred noise of the filter is 87 µVrms and the dynamic range is 58.1 dB. The filter offers the best figure of merit of 2.91 ×10−14 J, the lowest power consumption and voltage supply, compared with the previous state-of-the-art nanowatt filter designs. INDEX TERMS Biopotential filter, bulk-driven, low voltage, low power. I. INTRODUCTION In recent years, an increased interest in the design of ultralow-voltage (ULV) and ultra-low-power (ULP) electronic systems, devoted to biomedical applications, is observed. This new trend is associated with the development of different kind of implantable and wearable biomedical systems, where low dissipation power is of crucial importance [1], [2]. One of the most important blocks in such systems is the low-pass filter (LPF). As an example let us consider a typical electrocardiogram (ECG) acquisition system shown in Fig.1. The weak ECG signal, whose amplitude is in the range of 100 µV4 mV is first amplified by a low-noise preamplifier with variable gain, typically in the range of 10-100 V/V [1]. Next it is filtered with a LPF with adjustable cutoff frequency fc(100-250 Hz), to decrease outof-band noise. Finally, the signal is converted into digital form by an analog-to-digital converter (ADC) [1], [2]. Active filters devoted to such applications usually have very simple structures and exploit MOS transistors operating in sub-threshold region. This allows decreasing both, the dissipation power, as well as the occupied area. Simple structure allows decreasing the input referred noise as well. In recent years a number of ULV and ULP solutions for biopotential filters has been proposed [3]–[7]. In order to simplify their structures, usually the authors use a single The associate editor coordinating the review of this manuscript and approving it for publication was Dušan Grujić . FIGURE 1. ECG acquisition system. transistor as a transconductor, to realize a gm-C integrator. Their linearity and dynamic range (DR) is usually improved thanks to the local or global negative feedback loops. In order to further decrease their structures and current consumption, some of the proposed solutions of biquadratic sections contain only one branch of current, consisting of several stacked transistors [4]–[6]. However, the stacked transistors between the supply rails limit the minimum supply voltage (VDD) and entail different dc levels between input and output. In order to overcome the above mentioned constraints and further decrease both, the supply voltage, as well as the dissipation power (Pdiss) of an ECG filter, a bulk-driven (BD) technique can be considered. The BD circuits proved their capability to operate from extremely low VDD, even much below the threshold voltage (VTH) of MOS transistors, while maintaining the input common mode range (ICMR) almost rail-to-rail [7]–[13]. It is also worth noting, that in some cases the BD approach can extend the DR of analog circuit under ULV supply, since it can extend the input range for which 119586 This work is licensed under a Creative Commons Attribution 4.0 License. For more information, see https://creativecommons.org/licenses/by/4.0/ VOLUME 8, 2020 T. Kulej et al.: 0.3-V Nanopower Biopotential Low-Pass Filter FIGURE 2. CMOS schematic of the biquad filter (a) and its block diagram (b). the circuit performance is not limited by hard non-linearities. Therefore, the BD technique can be a promising approach to design ULV ECG filters. In this paper we propose a new solution for an ULV ECG filter based on the BD approach. The filter is a cascade connection of two newly-proposed ULV BD biquad filters. Similarly as in [4], the biquad was developed from a lowvoltage current re-use CMOS buffer [14]. Nevertheless, since the traditional long-tailed gate-driven (GD) differential pair was replaced with a non-tailed BD differential pair, a much lower minimum supply voltage was achieved. The lower VDD entails lower Pdiss of the overall structure. Moreover, the DR of the proposed biquad is better than reported for other designs in literature, even though its supply voltage is much lower (0.3 V). Due to the low dissipation power and good DR, the proposed filter outperforms other ECG filters in terms of standard Figures of Merit (FOMs). The rest of the paper is organized as follows. In Section II the newly-proposed BD biquad filter is discussed in detail. Section III describes the overall structure of the ECG filter. The simulation results and comparison with other similar designs are presented in Section IV. Finally, the paper is concluded in Section V. II. PROPOSED BIQUAD CELL A. CIRCUIT DESCRIPTION The schematic of the proposed ULV biquad filter is shown in Fig. 2. Its principle of operation is based on the idea of a simple CMOS unity-gain buffer [14]. This idea has been next adopted to signal filtering purposes, in a similar way as described in [4]. The circuit shown in Fig. 2. exploits similar operation principle, however, in this structure a traditional long-tail GD differential amplifier has been replaced by a non-tailed BD differential amplifier composed of the transistors M1A,B-M2A,B. Due to the non-tailed architecture combined with the BD approach, a rail-to-rail input swing can be achieved for very low supply voltage. The input differential amplifier, composed of the transistors M1A,B-M2A,B, exploits the idea first proposed in [15]. Let us assume that V1and V2represent the input voltages of the amplifier and the connection between the biquad output node and V2is broken. The transistors M1A-M2A and M1B-M2B form two current mirrors, biased with the current sinks IB1 = IB2 =IB. The input signals are applied to the bulk terminals of the transistors M1A,B-M2A,B, as shown in Fig. 2. Assuming V1=V2, the VBS voltages of all transistors are equal to each other, which entails that their threshold voltages are equal as well. Therefore, neglecting the impact of gds conductances, the currents I1and I2are also equal to each other and equal to IB. Consequently, for V1=V2, the differential output current of the first stage (I1-I2) is equal to zero, i.e. with the above assumptions, the input amplifier is insensitive to the commonmode voltage. For input differential signals, when V16=V2, the threshold voltages of M1A and M2A (M1B and M2B) are different, because of the dependence of the threshold voltages of MOS transistors on their bulk potentials. This affects the current transfer ratio of the current mirrors and produces a difference of the output currents I1and I2. Assuming that the p-channel transistor operates in saturated weak inversion region, its transfer characteristics can be approximated as: ID=IOW Lexp VSG +VTH npUT(1) where Iois the technology current, npis the subthreshold slope factor for p-channel MOS and UTis the thermal potential. The threshold voltage VTH, can be expressed as a function of VBS voltage as follows: VTH =VTHO −γpp2|φF|+VBS −p2|φF|(2) where VTHO is the threshold voltage for VBS =0, 2|8F| is twice the Fermi potential and γpis the bulk threshold parameter. Straightforward analysis shows that with the above model the differential output current of the first stage I1-I2can be expressed as: I1−I2= −2IBsinh ηV1−V2 npUT(3) VOLUME 8, 2020 119587 T. Kulej et al.: 0.3-V Nanopower Biopotential Low-Pass Filter where ηis the ratio of the bulk (gmbp) to gate (gmp) transconductance of the transistors M1A,B-M2A,Bat the operating point, given by: η=gmbp gmp =γp 2√2|φF|+VBS (4) As it can be concluded from the above considerations, the transistors M1A,B-M2A,Bform a truly differential (transconductance) amplifier, with non-linear large signal characteristic given by (3) and the small-signal transconductance which may be approximated as [15]: gm≈2gmbp (5) It is worth mentioning, that both, its noise properties and offset are not worse than observed for the corresponding BD differential pair biased with the same total current [10], [15]. The transistor M3B form a second transconductance stage. Consequently, the equivalent block diagram of the proposed biquad may be presented in the form shown in Fig.2.b where gmn =gm3B. It is worth pointing out, that due to the non-tailed architecture of the first transconductor in Fig. 2, the minimum supply voltage of the proposed biquad filter is as low as 2VDSsat, where VDSsat is the saturation voltage of an MOS transistor (3-4 UTin weak inversion region). The input/output swing is limited by the output characteristics of M1B and M3B. Because of the negative feedback loop, the maximum (minimum) output voltage can be very close to supply rails. B. TRANSFER FUNCTION The transfer function of the biquad filter in Fig. 2b can be expressed as follows: Vout (s) Vin(s)= gmbpgmn C1C2 s2+sgmn C2+gmbpgmn C1C2 (6) thus, it is transmittance of a low-pass filter, where its dc voltage gain (ADC), natural frequency (ωo), and quality factor (Q), are respectively given by: ADC =1 (7) ωo=rgmbpgmn C1C2 (8) Q=sgmbpC2 gmnC1 (9) Assuming gmn =IB/nnUTand gmbp =ηIB/npUT, the natural frequency and the quality factor may be expressed in the form: ωo=IB UTsη nnnpC1C2 (10) Q=sηnnC2 npC1 (11) As it is easy to note, the natural frequency of the biquad filter is proportional to the biasing current IBand can be easily TABLE 1. Transistors aspect ratios and filter capacitances. tuned with this current. On the other hand, variations of IB will not affect the quality factor Q. Both quantities (ωoand Q) will depend slightly on the process, supply voltage and temperature (PVT) variations, because the PVT variations will affect the slope factors nn, np, the coefficient ηand UT. However, one can expect that the variations will be on acceptable level. The tuning range of ωois limited by the acceptable range of |VGS/VDS|voltage drops across MOS transistors in this circuit, that provide operation of all transistors in saturation. C. NOISE PERFORMANCE The thermal v2 t,and flicker v2 1/f,noise spectral densities of a MOS transistor can be expressed as follows: v2 t=8kT 3gm (12) v2 1/f=Kg2 m fCoxWL (13) where k is the Boltzmann constant, T is the absolute temperature, gmis the transconductance, Cox is the gate oxide capacitance per unit area and K is the flicker noise constant. Assuming the above noise model, and neglecting the second order effects, the low frequency (f fc) input referred noise of the considered biquad in a weak inversion region can be expressed as: v2 tb =28kT 3ηgmbp 1+np nn(14) v2 1/fb =2 fCoxη2"Kp WL1,2+Kn WL3,5np nn2#(15) Usually, the thermal noise will be dominant, because of very low biasing currents (transconductances) and relatively large sizes of MOS transistors, required in ULV environment. Note, that the noise performance of the biquad filter will be deteriorated by the BD approach (η1), which is a well known disadvantage of all BD circuits. III. BIOPOTENTIAL FILTER The biopotential filter can be realized as a cascade connection of two biquad filters described in the previous section. Its transistor-level schematic is shown in Fig. 3. Transistor MB is used for biasing purposes. Note, that transistor sizes and biasing currents in both sections were assumed to be identical. 119588 VOLUME 8, 2020 T. Kulej et al.: 0.3-V Nanopower Biopotential Low-Pass Filter FIGURE 3. CMOS schematic of the proposed biopotential filter. FIGURE 4. AC response of the filter for IB =0.25nA, fc =100Hz. In order to obtain a fourth order maximally flat characteristic with 3-dB frequency of 100-Hz, for the assumed IB= 0.25 nA (gmbp =2.63 nS gmn =7.0 nS), the used capacitances were calculated as shown in Table 1. This results in fo/Q equal to 99.3 Hz/0.87 for the first and 129 Hz/0.65 for the second section of the filter respectively. The channel lengths of all transistors were chosen relatively large to maximize their gm/gds ratios (intrinsic voltage gains). On the other hand, their channel widths were fine tuned during the simulation phase to achieve |VGS| ≈VDD/2 at the operating point, that provides maximum voltage headroom for possible PVT variations and signal swing in an ULV environment. The transistor aspect ratios for the design are also provided in Table 1. IV. SIMULATED RESULTS A. MAIN CHARACTERISTICS The circuit has been implemented in a 0.18 µm CMOS process from TSMC, with threshold voltages of around +/− 0.5 V. Its performance was validated using Cadence/Spectre platform. The assumed supply voltage was 0.3 V (+/−0.15 V for the purpose of simulations). Fig. 4 shows the magnitude characteristic of the filter for IB=0.25 nA. The characteristic agree well with theory. The dc voltage gain was 0.144 dB, while the cutoff frequency was FIGURE 5. AC response of the filter for (IB =0.25nA, fc =100Hz), (IB = 0.377nA, fc =150Hz), (IB =0.634nA, fc =200Hz). FIGURE 6. Frequency fc versus bias current IB. exactly 100 Hz. The attenuation in the stop band exceeded 60 dB, which is sufficient for the considered application. Fig. 5 shows variations of the magnitude characteristic with the biasing current IB. For IBranging from 0.25 nA to 0.634 nA, the 3-dB frequency (fc) of the filter is tuned from 100 Hz to 250 Hz. Note, that variations of the dc voltage gain and shape of the characteristic (quality factors of both sections) are negligible, that agrees well with theory. Fig. 6. shows the cutoff frequency of the filter against the biasing current IB. As it is seen, the 3-dB frequency can be linearly tuned for IBranging from 0.15 nA to 0.65nA, which VOLUME 8, 2020 119589 T. Kulej et al.: 0.3-V Nanopower Biopotential Low-Pass Filter TABLE 2. Effect of PVT variation on the cutoff frequency. FIGURE 7. DC transfer characteristic of the filter showing rail-to-rail voltage range. FIGURE 8. Transient response of the filter for input sine wave with Vinpp =200mV and 10 Hz. THD =1%. corresponds to fcranging from 60 Hz to 260 Hz. This linear range is limited by the |VGS/VDS|voltage drops across MOS transistors and can be extended for larger VDD. Fig. 7 shows the static transfer characteristic of the filter. It is worth pointing out a very good linearity of this characteristic, for nearly rail-to-rail voltage swing. The large voltage swing in an ULV environment is obtained thanks to the BD approach applied in this design. The good linearity of the filter is confirmed also by its sine wave response shown in Fig. 8. For the sine wave of 200 mVpp and 10-Hz frequency, the total harmonic distortion (THD) was 1 %. The phase shift between input and output signals in Fig.8 is caused by the phase characteristic of the filter. The ability of the proposed filter to attenuate noise is shown in Fig. 9. In Fig. 9a we can see an input ECG signal with an amplitude of 50 mVp, interfered with a parasitic noise modeled as a 500-Hz sinusoidal signal with amplitude FIGURE 9. Transient response of the filter for noisy ECG signal, (a) input, (b) output. of 5mVp. Fig. 9b shows the resulting signal at the output of the filter, which confirms its proper operation in this case. B. IMPACT OF PVT VARIATIONS AND MISMATCH In order to investigate the filter sensitivity to PVT variations and transistor mismatch, both, the corner analysis as well as the Monte Carlo (MC) analysis have been performed. The results of corner analysis are shown in Table 2, where the impact of PVT variations on the cutoff frequency of the filter are shown and in Table 3, where its dc gain is examined. Both parameters show relatively low variations under the assumed range of PVT variations. The impact of transistor mismatch on fcand dc gain of the filter is shown in Figs. 10 and 11 respectively. The figures show histograms, being the results of MC analysis (200 runs). The observed standard deviations of fc(3.19 Hz) and dc gain (0.034 dB) prove a relatively low sensitivity of the filter to transistor mismatch. The input referred offset of the filter varied from −2.9 mV to 1.8 mV, as a result of PVT variations, while the MC 119590 VOLUME 8, 2020 T. Kulej et al.: 0.3-V Nanopower Biopotential Low-Pass Filter TABLE 3. Effect of PVT variation on the DC gain. TABLE 4. Performance comparison of nanopower 4th-order LP filters. FIGURE 10. The histogram of the frequency fc with 200 MC runs. mismatch analysis showed a mean value of −0.291 mV and standard deviation of 4.91 mV. The above results show, that the filter is robust under PVT variations and transistor mismatch. Note, that the impact of transistor mismatch could be further lowered, applying larger transistor channel sizes, at the cost of silicon area of the filter. C. COMPARISON Table 4 presents a comparison of the proposed filter with other similar designs of biopotential filters, published in FIGURE 11. The histogram of the DC gain with 200 MC runs. recent years. First of all is worth noting, that the proposed filter can operate with much lower VDD than other filters in Table 4, except [16]. Despite its single-ended architecture and ULV supply, the filter offers the best dynamic range among all the compared filters. This property is achieved thanks to the non-tailed BD architectures of the main blocks used in this design. Note, that the DR could be further improved with a fully-differential version of the proposed circuit. In order to facilitate the comparison, the following VOLUME 8, 2020 119591 T. Kulej et al.: 0.3-V Nanopower Biopotential Low-Pass Filter FIGURE 12. Comparison of FOMs and supply voltages. standard FOM has been used: FOM =Pdiss NfcDR (16) where N is the filter order and the other symbols were defined earlier. As it can be concluded from (16), the above FOM takes into account the power effectiveness of the filter, and refers it to the achieved DR. Lower value of this FOM means better performance of the filter. The values of the calculated FOMs are shown in Table 4 and in a graphical form in Fig. 12. As it is easy to note, the proposed filter offers the best FOM, which is approximately one order of magnitude better than achieved for the design in [16], which was supplied with the same VDD. The most similar FOM was reported for the filter in [5], however, that circuit was supplied with much larger VDD. V. CONCLUSION A new design for an ULV, fourth-order, biopotential (ECG) filter was presented. The filter is a cascade connection of two newly-proposed second-order (biquad) filters, developed from a compact current re-use CMOS buffers [4], [14]. Thanks to the application of BD non-tailed differential pairs, instead of traditional gate-driven pairs, a new ULV biquad filter was obtained, with very low supply voltage (0.3 V) and rail-to-rail input/output range. The simulated results showed, that the circuit outperforms all other designs in terms of standard FOMs, and operates well also in the presence of PVT variations and transistor mismatch. The filter performance can be further improved with a fully-differential version of this design. ACKNOWLEDGMENT For the research, infrastructure of the SIX Center was used. REFERENCES [1] S.-Y. Lee and C.-J. Cheng, ‘‘Systematic design and modeling of a OTAC filter for portable ECG detection,’’ IEEE Trans. Biomed. Circuits Syst., vol. 3, no. 1, pp. 53–64, Feb. 2009. [2] Y. Li, C. C. Y. Poon, and Y.-T. Zhang, ‘‘Analog integrated circuits design for processing physiological signals,’’ IEEE Rev. Biomed. Eng., vol. 3, pp. 93–105, Dec. 2010. [3] T. T. Zhang, P.-I. Mak, M.-I. Vai, P.-U. Mak, M.-K. Law, S.-H. Pun, and F. Wan, ‘‘15-nW biopotential LPFs in 0.35-µm CMOS using subthresholdsource-follower biquads with and without gain compensation,’’ IEEE Trans. Biomed. Circuits Syst., vol. 7, no. 5, pp. 690–702, Oct. 2013. [4] S. Thanapitak and C. Sawigun, ‘‘A subthreshold buffer-based biquadratic cell and its application to biopotential filter design,’’ IEEE Trans. Circuits Syst. I, Reg. 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Khateb, ‘‘Design and implementation of sub 0.5-V OTAs in 0.18-µm CMOS,’’ Int. J. Circuit Theory Appl., vol. 46, no. 6, pp. 1129–1143, Jun. 2018. [10] T. Kulej and F. Khateb, ‘‘A compact 0.3-V class AB bulk-driven OTA,’’ IEEE Trans. Very Large Scale Integr. (VLSI) Syst., vol. 28, no. 1, pp. 224–232, Jan. 2020, doi: 10.1109/TVLSI.2019.2937206. [11] F. Khateb and T. Kulej, ‘‘Design and implementation of a 0.3-V differential-difference amplifier,’’ IEEE Trans. Circuits Syst. I, Reg. Papers, vol. 66, no. 2, pp. 513–523, Feb. 2019. [12] T. Kulej and F. Khateb, ‘‘A 0.3-V 98-dB rail-to-rail OTA in 0.18 µmCMOS,’’ IEEE Access, vol. 8:, pp. 27459–27467, 2020. [13] F. Khateb, T. Kulej, and M. Kumngern, ‘‘0.3 V bulk-driven current conveyor,’’ IEEE Access, vol. 7, pp. 65122–65128, 2019. [14] G. Palmisano, G. Palumbo, and S. Pennisi, ‘‘High-performance and simple CMOS unity-gain amplifier,’’ IEEE Trans. Circuits Syst. I, Fundam. Theory Appl., vol. 47, no. 3, pp. 406–410, Mar. 2000. [15] T. Kulej, ‘‘0.5-V bulk-driven CMOS operational amplifier,’’ IET Circuits, Devices Syst., vol. 7, no. 6, pp. 352–360, Nov. 2013. [16] P. Sreenivasulu, G. Hanumantha Rao, S. Rekha, and M. S. Bhat, ‘‘A 0.3 V, 56 dB DR, 100 hz fourth order low-pass filter for ECG acquisition system,’’ Microelectron. J., vol. 94, Dec. 2019, Art. no. 104652. TOMASZ KULEJ received the M.Sc. and Ph.D. degrees (Hons.) from the Gdańsk University of Technology, Gdańsk, Poland, in 1990 and 1996, respectively. He was a Senior Design Analysis Engineer with the Polish Branch, Chipworks Inc., Ottawa, ON, Canada. He is currently an Associate Professor with the Department of Electrical Engineering, Częstochowa University of Technology, Poland, where he conducts lectures on electronics fundamentals, analog circuits, and computer aided design. He has authored or coauthored over 70 publications in peer-reviewed journals and conferences. He holds three patents. His recent research interests include analog integrated circuits in CMOS technology, with emphasis to low-voltage and low-power solutions. He serves as an Associate Editor for Circuits, Systems, and Signal Processing and IET Circuits, Devices and Systems. He served as a Guest Editor for the Special Issues on Low Voltage Integrated Circuits on Circuits, Systems, and Signal Processing (2017), IET Circuits Devices and Systems (2018), and Microelectronics Journal (2019). 119592 VOLUME 8, 2020 T. Kulej et al.: 0.3-V Nanopower Biopotential Low-Pass Filter FABIAN KHATEB received the M.Sc. and Ph.D. degrees in electrical engineering and communication and business and management from the Brno University of Technology, Czech Republic, in 2002, 2003, 2005, and 2007, respectively. He is currently a Professor with the Department of Microelectronics, Faculty of Electrical Engineering and Communication, Brno University of Technology. He is also with the Department of Information and Communication Technology in Medicine, Faculty of Biomedical Engineering, Czech Technical University in Prague. He holds five patents. He has authored or coauthored over 100 publications in journals and proceedings of international conferences. He has expertise in new principles of designing low-voltage low-power analog circuits, particularly biomedical applications. He is a member of the Editorial Board of Microelectronics Journal. He serves as an Associate Editor for Circuits, Systems and Signal Processing,IET Circuits, Devices and Systems, and the International Journal of Electronics. He was a Guest Editor for the Special Issue on Current-Mode Circuits and Systems, Recent Advances, Design and Applications on International Journal of Electronics and Communications (2017). He was also a Lead Guest Editor for the Special Issues on Low Voltage Integrated Circuits and Systems on Circuits, Systems, and Signal Processing (2017), IET Circuits, Devices and Systems (2018), and Microelectronics Journal (2019). MONTREE KUMNGERN received the B.S.Ind.Ed. degree in electrical engineering from the King Mongkut’s University of Technology Thonburi, Thailand, in 1998, and the M.Eng. and D.Eng. degrees in electrical engineering from the King Mongkut’s Institute of Technology Ladkrabang, Thailand, in 2002 and 2006, respectively. From 2007, he served as a Lecturer with the Department of Telecommunications Engineering, Faculty of Engineering, King Mongkut’s Institute of Technology Ladkrabang. From 2010 to 2017, he served as an Assistant Professor and he is currently an Associate Professor. He has authored or coauthored over 200 publications in journals and proceedings of international conferences. His research interests include analog and digital integrated circuits, discretetime analog filters, non-linear circuits, data converters, and ultra-low voltage building blocks for biomedical applications. VOLUME 8, 2020 119593